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Intersil CA3140

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Intersil CA3140CA3140EZ CA3140 4.5MHZ BiMOS OP-AMP IC
Op Amp Type:High Speed; No. of Amplifiers:1; Bandwidth:4.5MHz; No. of Pins:8; Operating Temperature Range:-55°C to +125°C ; RoHS Compliant: Yes

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Part Number: CA3140EZ


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FN957.10 July 11, 2005

Absolute Maximum Ratings
DC Supply Voltage (Between V+ and V- Terminals). 36V Differential Mode Input Voltage. 8V DC Input Voltage. (V+ +8V) To (V- -0.5V) Input Terminal Current. 1mA Output Short Circuit Duration (Note 2). Indefinite Operating Conditions Temperature Range. -55oC to 125oC

Thermal Information

Thermal Resistance (Typical, Note 1) JA (oC/W) JC (oC/W) PDIP Package*. 115 N/A SOIC Package. 165 N/A Maximum Junction Temperature (Plastic Package). 150oC Maximum Storage Temperature Range. -65oC to 150oC Maximum Lead Temperature (Soldering 10s). 300oC (SOIC - Lead Tips Only) *Pb-free PDIPs can be used for through hole wave solder processing only. They are not intended for use in Reflow solder processing applications.
CAUTION: Stresses above those listed in Absolute Maximum Ratings may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTES: 1. JA is measured with the component mounted on a low effective thermal conductivity test board in free air. See Tech Brief TB379 for details 2. Short circuit may be applied to ground or to either supply.
Electrical Specifications
VSUPPLY = 15V, TA = 25oC TYPICAL VALUES
PARAMETER Input Offset Voltage Adjustment Resistor

SYMBOL

TEST CONDITIONS Typical Value of Resistor Between Terminals 4 and 5 or 4 and 1 to Adjust Max VIO

CA3140 4.7

CA3140A 18

UNITS k

Input Resistance Input Capacitance Output Resistance Equivalent Wideband Input Noise Voltage (See Figure 27) Equivalent Input Noise Voltage (See Figure 35)
RI CI RO eN eN BW = 140kHz, RS = 1M RS = 100 f = 1kHz f = 10kHz

1.4.220

1.4.220 0.4.5 1.4
T pF V nV/Hz nV/Hz mA mA MHz V/s A s % s s
Short Circuit Current to Opposite Supply

IOM+ IOM-

Source Sink
Gain-Bandwidth Product, (See Figures 6, 30) Slew Rate, (See Figure 31) Sink Current From Terminal 8 To Terminal 4 to Swing Output Low Transient Response (See Figure 28)
RL = 2k CL = 100pF RL = 2k CL = 100pF Voltage Follower
Rise Time Overshoot To 1mV To 10mV

0.4.5 1.4

Settling Time at 10VP-P, (See Figure 5)
PARAMETER Input Offset Voltage Input Offset Current Input Current
For Equipment Design, at VSUPPLY = 15V, TA = 25oC, Unless Otherwise Specified CA3140 SYMBOL |VIO| |IIO| II MIN TYP 5 0.MAX 50 MIN CA3140A TYP 2 0.MAX 40 UNITS mV pA pA

PARAMETER Large Signal Voltage Gain (Note 3) (See Figures 6, 29) Common Mode Rejection Ratio (See Figure 34) Common Mode Input Voltage Range (See Figure 8) Power-Supply Rejection Ratio, VIO/VS (See Figure 36) Max Output Voltage (Note 4) (See Figures 2, 8) Supply Current (See Figure 32) Device Dissipation Input Offset Voltage Temperature Drift NOTES: 3. At VO = 26VP-P , +12V, -14V and RL = 2k. 4. At RL = 2k. For Equipment Design, at VSUPPLY = 15V, TA = 25oC, Unless Otherwise Specified (Continued) CA3140 SYMBOL AOL MIN CMRR 70 VICR PSRR -VOM+ VOMI+ PD VIO/T +12 -14 TYP -15.5 to +12.-14.MAX 180 MIN 70 -+12 -14 CA3140A TYP -15.5 to +12.-14.MAX 180 UNITS kV/V dB V/V dB V V/V dB V V mA mW V/oC
For Design Guidance At V+ = 5V, V- = 0V, TA = 25oC TYPICAL VALUES
PARAMETER Input Offset Voltage Input Offset Current Input Current Input Resistance Large Signal Voltage Gain (See Figures 6, 29)
SYMBOL |VIO| |IIO| II RI AOL

CA0.100

CA3140A 2 0.90 -0.5 2.0.3.7 1.200
UNITS mV pA pA T kV/V dB V/V dB V V V/V dB V V mA mA V/s MHz mA mW A
Common Mode Rejection Ratio
Common Mode Input Voltage Range (See Figure 8)

-0.5 2.6

Power Supply Rejection Ratio

PSRR VIO/VS VOM+ VOM-

3 0.3.7 1.200
Maximum Output Voltage (See Figures 2, 8)

Maximum Output Current:

IOM+ I
Slew Rate (See Figure 31) Gain-Bandwidth Product (See Figure 30) Supply Current (See Figure 32) Device Dissipation Sink Current from Terminal 8 to Terminal 4 to Swing Output Low

SR fT I+ PD

CA3140, CA3140A Block Diagram
2mA BIAS CIRCUIT CURRENT SOURCES AND REGULATOR + 3 INPUT 200A 1.6mA 200A A 10,000 C1 12pF STROBE 4 V8 2A 2mA 4mA 7 V+

6 OUTPUT

OFFSET NULL

Schematic Diagram

BIAS CIRCUIT INPUT STAGE SECOND STAGE OUTPUT STAGE DYNAMIC CURRENT SINK 7 V+ D1 Q3 D7 Q2 RR10 1K Q19 RR12 12K Q20 R13 5K

D8 R14 20K

Q17 R1 8K Q8 R8 1K Q 18
6 OUTPUT D2 D3 D4 D5 INVERTING INPUT NON-INVERTING INPUT 2 + 3 RR3 500
Q10 C1 12pF Q14 Q15 Q16 D6 RRQ13

R4 500

R5 500

5 OFFSET NULL

8 STROBE
All resistance values are in ohms.
CA3140, CA3140A Application Information

Circuit Description

As shown in the block diagram, the input terminals may be operated down to 0.5V below the negative supply rail. Two class A amplifier stages provide the voltage gain, and a unique class AB amplifier stage provides the current gain necessary to drive low-impedance loads. A biasing circuit provides control of cascoded constant current flow circuits in the first and second stages. The CA3140 includes an on chip phase compensating capacitor that is sufficient for the unity gain voltage follower configuration. When the CA3140 is operating such that output Terminal 6 is sinking current to the V- bus, transistor Q16 is the current sinking element. Transistor Q16 is mirror connected to D6, R7, with current fed by way of Q21, R12, and Q20. Transistor Q20, in turn, is biased by current flow through R13, zener D8, and R14. The dynamic current sink is controlled by voltage level sensing. For purposes of explanation, it is assumed that output Terminal 6 is quiescently established at the potential midpoint between the V+ and V- supply rails. When output current sinking mode operation is required, the collector potential of transistor Q13 is driven below its quiescent level, thereby causing Q17, Q18 to decrease the output voltage at Terminal 6. Thus, the gate terminal of PMOS transistor Q21 is displaced toward the V- bus, thereby reducing the channel resistance of Q21. As a consequence, there is an incremental increase in current flow through Q20, R12, Q21, D6, R7, and the base of Q16. As a result, Q16 sinks current from Terminal 6 in direct response to the incremental change in output voltage caused by Q18. This sink current flows regardless of load; any excess current is internally supplied by the emitter-follower Q18. Short circuit protection of the output circuit is provided by Q19, which is driven into conduction by the high voltage drop developed across R11 under output short circuit conditions. Under these conditions, the collector of Q19 diverts current from Q4 so as to reduce the base current drive from Q17, thereby limiting current flow in Q18 to the short circuited load terminal.

Input Stage

The schematic diagram consists of a differential input stage using PMOS field-effect transistors (Q9, Q10) working into a mirror pair of bipolar transistors (Q11, Q12) functioning as load resistors together with resistors R2 through R5. The mirror pair transistors also function as a differential-to-single-ended converter to provide base current drive to the second stage bipolar transistor (Q13). Offset nulling, when desired, can be effected with a 10k potentiometer connected across Terminals 1 and 5 and with its slider arm connected to Terminal 4. Cascode-connected bipolar transistors Q2, Q5 are the constant current source for the input stage. The base biasing circuit for the constant current source is described subsequently. The small diodes D3, D4, D5 provide gate oxide protection against high voltage transients, e.g., static electricity.

Bias Circuit

Quiescent current in all stages (except the dynamic current sink) of the CA3140 is dependent upon bias current flow in R1. The function of the bias circuit is to establish and maintain constant current flow through D1, Q6, Q8 and D2. D1 is a diode connected transistor mirror connected in parallel with the base emitter junctions of Q1, Q2, and Q3. D1 may be considered as a current sampling diode that senses the emitter current of Q6 and automatically adjusts the base current of Q6 (via Q1) to maintain a constant current through Q6, Q8, D2. The base currents in Q2, Q3 are also determined by constant current flow D1. Furthermore, current in diode connected transistor Q2 establishes the currents in transistors Q14 and Q15.

Second Stage

Most of the voltage gain in the CA3140 is provided by the second amplifier stage, consisting of bipolar transistor Q13 and its cascode connected load resistance provided by bipolar transistors Q3, Q4. On-chip phase compensation, sufficient for a majority of the applications is provided by C1. Additional Miller-Effect compensation (roll off) can be accomplished, when desired, by simply connecting a small capacitor between Terminals 1 and 8. Terminal 8 is also used to strobe the output stage into quiescence. When terminal 8 is tied to the negative supply rail (Terminal 4) by mechanical or electrical means, the output Terminal 6 swings low, i.e., approximately to Terminal 4 potential.

Typical Applications

Wide dynamic range of input and output characteristics with the most desirable high input impedance characteristics is achieved in the CA3140 by the use of an unique design based upon the PMOS Bipolar process. Input common mode voltage range and output swing capabilities are complementary, allowing operation with the single supply down to 4V. The wide dynamic range of these parameters also means that this device is suitable for many single supply applications, such as, for example, where one input is driven below the potential of Terminal 4 and the phase sense of the output signal must be maintained a most important consideration in comparator applications.

Output Stage

The CA3140 Series circuits employ a broad band output stage that can sink loads to the negative supply to complement the capability of the PMOS input stage when operating near the negative rail. Quiescent current in the emitter-follower cascade circuit (Q17, Q18) is established by transistors (Q14, Q15) whose base currents are mirrored to current flowing through diode D2 in the bias circuit section. When the CA3140 is operating such that output Terminal 6 is sourcing current, transistor Q18 functions as an emitter-follower to source current from the V+ bus (Terminal 7), via D7, R9, and R11. Under these conditions, the collector potential of Q13 is sufficiently high to permit the necessary flow of base current to emitter follower Q17 which, in turn, drives Q18. 6
Output Circuit Considerations
Excellent interfacing with TTL circuitry is easily achieved with a single 6.2V zener diode connected to Terminal 8 as shown in Figure 1. This connection assures that the maximum output signal swing will not go more positive than the zener voltage minus two base-to-emitter voltage drops within the CA3140. These voltages are independent of the operating supply voltage.
V+ 5V TO 36V 8 CA4 6.2V 6 LOGIC SUPPLY 5V TYPICAL TTL GATE
level shifting circuitry usually associated with the 741 series of operational amplifiers. Figure 4 shows some typical configurations. Note that a series resistor, RL, is used in both cases to limit the drive available to the driven device. Moreover, it is recommended that a series diode and shunt diode be used at the thyristor input to prevent large negative transient surges that can appear at the gate of thyristors, from damaging the integrated circuit.

Offset Voltage Nulling

The input offset voltage can be nulled by connecting a 10k potentiometer between Terminals 1 and 5 and returning its wiper arm to terminal 4, see Figure 3A. This technique, however, gives more adjustment range than required and therefore, a considerable portion of the potentiometer rotation is not fully utilized. Typical values of series resistors (R) that may be placed at either end of the potentiometer, see Figure 3B, to optimize its utilization range are given in the Electrical Specifications table. An alternate system is shown in Figure 3C. This circuit uses only one additional resistor of approximately the value shown in the table. For potentiometers, in which the resistance does not drop to 0 at either end of rotation, a value of resistance 10% lower than the values shown in the table should be used.

For those cases where bandwidth reduction is desired, for example, broadband noise reduction, an external capacitor connected between Terminals 1 and 8 can reduce the open loop -3dB bandwidth. The slew rate will, however, also be proportionally reduced by using this additional capacitor. Thus, a 20% reduction in bandwidth by this technique will also reduce the slew rate by about 20%. Figure 5 shows the typical settling time required to reach 1mV or 10mV of the final value for various levels of large signal inputs for the voltage follower and inverting unity gain amplifiers. 8
The exceptionally fast settling time characteristics are largely due to the high combination of high gain and wide bandwidth of the CA3140; as shown in Figure 6.
Input Circuit Considerations
As mentioned previously, the amplifier inputs can be driven below the Terminal 4 potential, but a series current limiting resistor is recommended to limit the maximum input terminal current to less than 1mA to prevent damage to the input protection circuitry. Moreover, some current limiting resistance should be provided between the inverting input and the output when
the CA3140 is used as a unity gain voltage follower. This resistance prevents the possibility of extremely large input signal transients from forcing a signal through the input protection network and directly driving the internal constant current source which could result in positive feedback via the output terminal. A 3.9k resistor is sufficient. The typical input current is on the order of 10pA when the inputs are centered at nominal device dissipation. As the output supplies load current, device dissipation will increase, raising the chip temperature and resulting in increased input current. Figure 7 shows typical input terminal current versus ambient temperature for the CA3140. It is well known that MOSFET devices can exhibit slight changes in characteristics (for example, small changes in
OPEN LOOP PHASE (DEGREES)
input offset voltage) due to the application of large differential input voltages that are sustained over long periods at elevated temperatures. Both applied voltage and temperature accelerate these changes. The process is reversible and offset voltage shifts of the opposite polarity reverse the offset. Figure 9 shows the typical offset voltage change as a function of various stress voltages at the maximum rating of 125oC (for metal can); at lower temperatures (metal can and plastic), for example, at 85oC, this change in voltage is considerably less. In typical linear applications, where the differential voltage is small and symmetrical, these incremental changes are of about the same magnitude as those encountered in an operational amplifier employing a bipolar transistor input stage.

OPEN LOOP VOLTAGE GAIN (dB)
SUPPLY VOLTAGE: VS = 15V TA = 25oC 101 RL = 2k, CL = 100pF OL
-75 RL = 2k, CL = 0pF -90 -105 -120 -135 -150

SUPPLY VOLTAGE: VS = 15V

INPUT CURRENT (pA)

106 FREQUENCY (Hz)

TEMPERATURE (oC)
FIGURE 6. OPEN LOOP VOLTAGE GAIN AND PHASE vs FREQUENCY
FIGURE 7. INPUT CURRENT vs TEMPERATURE
INPUT AND OUTPUT VOLTAGE EXCURSIONS FROM TERMINAL 7 (V+)
RL = 0 -0.5 -1.0 -1.5 -2.0 -2.5 -3.0 +VICR AT TA = 125oC +VICR AT TA = 25oC +VICR AT TA = -55oC +VOUT AT TA = 125oC +VOUT AT TA = 25oC +VOUT AT TA = -55oC
INPUT AND OUTPUT VOLTAGE EXCURSIONS FROM TERMINAL 4 (V-)
1.5 1.0 0.-0.5 -1.0 -1.5 -VOUT FOR TA = -55oC to 125oC -VICR AT TA = 125oC -VICR AT TA = 25oC -VICR AT TA = -55oC

SUPPLY VOLTAGE (V+, V-)

FIGURE 8. OUTPUT VOLTAGE SWING CAPABILITY AND COMMON MODE INPUT VOLTAGE RANGE vs SUPPLY VOLTAGE
7 OFFSET VOLTAGE SHIFT (mV) 4500 TIME (HOURS) DIFFERENTIAL DC VOLTAGE (ACROSS TERMINALS 2 AND 3) = 0V OUTPUT VOLTAGE = V+ / 2 TA = 125oC FOR METAL CAN PACKAGES DIFFERENTIAL DC VOLTAGE (ACROSS TERMINALS 2 AND 3) = 2V OUTPUT STAGE TOGGLED
placed across the input to the CA3080A to give a logarithmic analog indication of the function generators frequency. Analog frequency readout is readily accomplished by the means described above because the output current of the CA3080A varies approximately one decade for each 60mV change in the applied voltage, VABC (voltage between Terminals 5 and 4 of the CA3080A of the function generator). Therefore, six decades represent 360mV change in VABC. Now, only the reference voltage must be established to set the lower limit on the meter. The three remaining transistors from the CA3086 Array used in the sweep generator are used for this reference voltage. In addition, this reference generator arrangement tends to track ambient temperature variations, and thus compensates for the effects of the normal negative temperature coefficient of the CA3080A VABC terminal voltage. Another output voltage from the reference generator is used to insure temperature tracking of the lower end of the Frequency Adjustment Potentiometer. A large series resistance simulates a current source, assuring similar temperature coefficients at both ends of the Frequency Adjustment Control. To calibrate this circuit, set the Frequency Adjustment Potentiometer at its low end. Then adjust the Minimum Frequency Calibration Control for the lowest frequency. To establish the upper frequency limit, set the Frequency Adjustment Potentiometer to its upper end and then adjust the Maximum Frequency Calibration Control for the maximum frequency. Because there is interaction among these controls, repetition of the adjustment procedure may be necessary. Two adjustments are used for the meter. The meter sensitivity control sets the meter scale width of each decade, while the meter position control adjusts the pointer on the scale with negligible effect on the sensitivity adjustment. Thus, the meter sensitivity adjustment control calibrates the meter so that it deflects 1/6 of full scale for each decade change in frequency.

FIGURE 9. TYPICAL INCREMENTAL OFFSET VOLTAGE SHIFT vs OPERATING LIFE
Super Sweep Function Generator
A function generator having a wide tuning range is shown in Figure 10. The 1,000,000/1 adjustment range is accomplished by a single variable potentiometer or by an auxiliary sweeping signal. The CA3140 functions as a noninverting readout amplifier of the triangular signal developed across the integrating capacitor network connected to the output of the CA3080A current source. Buffered triangular output signals are then applied to a second CA3080 functioning as a high speed hysteresis switch. Output from the switch is returned directly back to the input of the CA3080A current source, thereby, completing the positive feedback loop The triangular output level is determined by the four 1N914 level limiting diodes of the second CA3080 and the resistor divider network connected to Terminal No. 2 (input) of the CA3080. These diodes establish the input trip level to this switching stage and, therefore, indirectly determine the amplitude of the output triangle. Compensation for propagation delays around the entire loop is provided by one adjustment on the input of the CA3080. This adjustment, which provides for a constant generator amplitude output, is most easily made while the generator is sweeping. High frequency ramp linearity is adjusted by the single 7pF to 60pF capacitor in the output of the CA3080A. It must be emphasized that only the CA3080A is characterized for maximum output linearity in the current generator function.

Sine Wave Shaper

The circuit shown in Figure 12 uses a CA3140 as a voltage follower in combination with diodes from the CA3019 Array to convert the triangular signal from the function generator to a sine-wave output signal having typically less than 2% THD. The basic zero crossing slope is established by the 10k potentiometer connected between Terminals 2 and 6 of the CA3140 and the 9.1k resistor and 10k potentiometer from Terminal 2 to ground. Two break points are established by diodes D1 through D4. Positive feedback via D5 and D6 establishes the zero slope at the maximum and minimum levels of the sine wave. This technique is necessary because the voltage follower configuration approaches unity gain rather than the zero gain required to shape the sine wave at the two extremes.
Meter Driver and Buffer Amplifier
Figure 11 shows the CA3140 connected as a meter driver and buffer amplifier. Low driving impedance is required of the CA3080A current source to assure smooth operation of the Frequency Adjustment Control. This low-driving impedance requirement is easily met by using a CA3140 connected as a voltage follower. Moreover, a meter may be

51k +15V

CA3140 + 4

10k GATE PULSE OUTPUT

EXTERNAL OUTPUT
TO FUNCTION GENERATOR SWEEP IN SWEEP WIDTH +15V

LOGVIO

4 51k 6.8k 91k

1 25k 3.9 -15V 5

10k TRIANGLE
1 TRANSISTORS FROM CA3086 ARRAY

SAWTOOTH LOG

100 390
FIGURE 13. SWEEPING GENERATOR
This circuit can be adjusted most easily with a distortion analyzer, but a good first approximation can be made by comparing the output signal with that of a sine wave generator. The initial slope is adjusted with the potentiometer R1, followed by an adjustment of R2. The final slope is established by adjusting R3, thereby adding additional segments that are contributed by these diodes. Because there is some interaction among these controls, repetition of the adjustment procedure may be necessary. Sweeping Generator Figure 13 shows a sweeping generator. Three CA3140s are used in this circuit. One CA3140 is used as an integrator, a second device is used as a hysteresis switch that determines the starting and stopping points of the sweep. A third CA3140 is used as a logarithmic shaping network for the log function. Rates and slopes, as well as sawtooth, triangle, and logarithmic sweeps are generated by this circuit. Wideband Output Amplifier Figure 14 shows a high slew rate, wideband amplifier suitable for use as a 50 transmission line driver. This circuit, when used in conjunction with the function generator and sine wave shaper circuits shown in Figures 10 and 12 provides 18VP-P output open circuited, or 9VP-P output when terminated in 50. The slew rate required of this amplifier is 28V/s (18VP-P x x 0.5MHz).
+15V SIGNAL LEVEL ADJUSTMENT 2.5k 2

REFERENCE VOLTAGE INPUT

VOLTAGE ADJUSTMENT 3

REGULATED OUTPUT

CA3140 2
FIGURE 15. BASIC SINGLE SUPPLY VOLTAGE REGULATOR SHOWING VOLTAGE FOLLOWER CONFIGURATION
Essentially, the regulators, shown in Figures 16 and 17, are connected as non inverting power operational amplifiers with a gain of 3.2. An 8V reference input yields a maximum output voltage slightly greater than 25V. As a voltage follower, when the reference input goes to 0V the output will be 0V. Because the offset voltage is also multiplied by the 3.2 gain factor, a potentiometer is needed to null the offset voltage. Series pass transistors with high ICBO levels will also prevent the output voltage from reaching zero because there is a finite voltage drop (VCESAT) across the output of the CA3140 (see Figure 2). This saturation voltage level may indeed set the lowest voltage obtainable. The high impedance presented by Terminal 8 is advantageous in effecting current limiting. Thus, only a small signal transistor is required for the current-limit sensing amplifier. Resistive decoupling is provided for this transistor to minimize damage to it or the CA3140 in the event of unusual input or output transients on the supply rail. Figures 16 and 17, show circuits in which a D2201 high speed diode is used for the current sensor. This diode was chosen for its slightly higher forward voltage drop characteristic, thus giving greater sensitivity. It must be emphasized that heat sinking of this diode is essential to minimize variation of the current trip point due to internal heating of the diode. That is, 1A at 1V forward drop represents one watt which can result in significant regenerative changes in the current trip point as the diode temperature rises. Placing the small signal reference amplifier in the proximity of the current sensing diode also helps minimize the variability in the trip level due to the negative temperature coefficient of the diode. In spite of those limitations, the current limiting point can easily be adjusted over the range from 10mA to 1A with a single adjustment potentiometer. If the temperature stability of the current limiting system is a serious consideration, the more usual current sampling resistor type of circuitry should be employed. A power Darlington transistor (in a metal can with heatsink), is used as the series pass element for the conventional current limiting system, Figure 16, because high power Darlington dissipation will be encountered at low output voltage and high currents.

50F 25V

2N3053

1N1N914

2.7 2.7
OUTPUT DC LEVEL ADJUSTMENT

+15V 3k -15V 200

2.4pF 2pF

2N4037 -15V

NOMINAL BANDWIDTH = 10MHz tr = 35ns
FIGURE 14. WIDEBAND OUTPUT AMPLIFIER
Power Supplies High input impedance, common mode capability down to the negative supply and high output drive current capability are key factors in the design of wide range output voltage supplies that use a single input voltage to provide a regulated output voltage that can be adjusted from essentially 0V to 24V. Unlike many regulator systems using comparators having a bipolar transistor input stage, a high impedance reference voltage divider from a single supply can be used in connection with the CA3140 (see Figure 15).
A small heat sink VERSAWATT transistor is used as the series pass element in the fold back current system, Figure 17, since dissipation levels will only approach 10W. In this system, the D2201 diode is used for current sampling. Foldback is provided by the 3k and 100k divider network connected to the base of the current sensing transistor. Both regulators provide better than 0.02% load regulation. Because there is constant loop gain at all voltage settings, the
2N6385 CURRENT POWER DARLINGTON LIMITING ADJUST D75 3k 6 2.7k 10F INPUT 2.2k 12 0.01F 4 CA3086 1k 1k 62k HUM AND NOISE OUTPUT <200VRMS (MEASUREMENT BANDWIDTH ~10MHz) LINE REGULATION 0.1%/V LOAD REGULATION (NO LOAD TO FULL LOAD) <0.02% 62k HUM AND NOISE OUTPUT <200VRMS (MEASUREMENT BANDWIDTH ~10MHz) LINE REGULATION 0.1%/V LOAD REGULATION (NO LOAD TO FULL LOAD) <0.02% 4 CA3086 + + CA100k VOLTAGE ADJUST 100k + 3 INPUT 2.2k 0.01F 13 + 1k 82k 2.7k 10F + 8 1k 56pF 180k 2 CA100k VOLTAGE ADJUST 100k + 3 1k 82k 1k 1k 1k 2 2N2102 3
regulation also remains constant. Line regulation is 0.1% per volt. Hum and noise voltage is less than 200V as read with a meter having a 10MHz bandwidth. Figure 18A shows the turn ON and turn OFF characteristics of both regulators. The slow turn on rise is due to the slow rate of rise of the reference voltage. Figure 18B shows the transient response of the regulator with the switching of a 20 load at 20V output.
FOLDBACK CURRENT LIMITER 2N5294 D3 1k 200 OUTPUT 0V TO 25V 25V AT 1A FOLDS BACK TO 40mA

OUTPUT 0.1 24V AT 1A 1

1 100k

3k 2N1k 56pF 180k

5F 50k 14
FIGURE 16. REGULATED POWER SUPPLY
FIGURE 17. REGULATED POWER SUPPLY WITH FOLDBACK CURRENT LIMITING

5V/Div., 1s/Div.

Top Trace: Output Voltage; 200mV/Div., 5s/Div. Bottom Trace: Collector of load switching transistor, load = 1A; 5V/Div., 5s/Div. FIGURE 18B. TRANSIENT RESPONSE
FIGURE 18A. SUPPLY TURN-ON AND TURNOFF CHARACTERISTICS
FIGURE 18. WAVEFORMS OF DYNAMIC CHARACTERISTICS OF POWER SUPPLY CURRENTS SHOWN IN FIGURES 16 AND 17

Tone Control Circuits

High slew rate, wide bandwidth, high output voltage capability and high input impedance are all characteristics required of tone control amplifiers. Two tone control circuits that exploit these characteristics of the CA3140 are shown in Figures 19 and 20. The first circuit, shown in Figure 20, is the Baxandall tone control circuit which provides unity gain at midband and uses standard linear potentiometers. The high input impedance of the CA3140 makes possible the use of lowcost, low-value, small size capacitors, as well as reduced load of the driving stage. Bass treble boost and cut are 15dB at 100Hz and 10kHz, respectively. Full peak-to-peak output is available up to at least 20kHz due to the high slew rate of the CA3140. The amplifier gain is 3dB down from its flat position at 70kHz. Figure 19 shows another tone control circuit with similar boost and cut specifications. The wideband gain of this circuit is equal to the ultimate boost or cut plus one, which in this case is a gain of eleven. For 20dB boost and cut, the input loading of this circuit is essentially equal to the value of the resistance from Terminal No. 3 to ground. A detailed analysis of this circuit is given in An IC Operational Transconductance Amplifier (OTA) With Power Capability by L. Kaplan and H. Wittlinger, IEEE Transactions on Broadcast and Television Receivers, Vol. BTR-18, No. 3, August, 1972.
NOTES: 5. 20dB Flat Position Gain.

5.1 M 2 + CA3140 0.1F 6

FOR SINGLE SUPPLY +30V 2.2M 0.005F
6. 15dB Bass and Treble Boost and Cut at 100Hz and 10kHz, respectively. 7. 25VP-P output at 20kHz. 8. -3dB at 24kHz from 1kHz reference.
FOR DUAL SUPPLIES +15V 0.005F 3 5.1M + CA3140 0.1F 6
BOOST 0.012F 2.2M 0.1 F 18k
TREBLE CUT 200k (LINEAR) 0.001F 100 pF 100pF

4 0.1F -15V

0.022F 2F - + 10k

0.0022F

1M 100k CCW (LOG) BOOST BASS CUT TONE CONTROL NETWORK

TONE CONTROL NETWORK

FIGURE 19. TONE CONTROL CIRCUIT USING CA3130 SERIES (20dB MIDBAND GAIN)
FOR SINGLE SUPPLY BOOST 0.047F BASS CUT (LINEAR) 240k 240k 5M 2.2M 750 pF 750 pF 3 2.2M 0.1 F 2.2 M 2 +32V +15V 7 + CA3140 0.1 F 6 0.047F 4 20pF 5M 51k (LINEAR) BOOST TREBLE CUT TONE CONTROL NETWORK 51k TONE CONTROL NETWORK 7 + CA3140 0.1F 6

FOR DUAL SUPPLIES

4 -15V 0.1F
: 9. 15dB Bass and Treble Boost and Cut at 100Hz and 10kHz, Respectively. 10. 25VP-P Output at 20kHz. 11. -3dB at 70kHz from 1kHz Reference. 12. 0dB Flat Position Gain.
FIGURE 20. BAXANDALL TONE CONTROL CIRCUIT USING CA3140 SERIES

Wien Bridge Oscillator

Another application of the CA3140 that makes excellent use of its high input impedance, high slew rate, and high voltage qualities is the Wien Bridge sine wave oscillator. A basic Wien Bridge oscillator is shown in Figure 21. When R1 = R2 = R and C1 = C2 = C, the frequency equation reduces to the familiar f = 1/(2RC) and the gain required for oscillation, AOSC is equal to 3. Note that if C2 is increased by a factor of four and R2 is reduced by a factor of four, the gain required for oscillation becomes 1.5, thus permitting a potentially higher operating frequency closer to the gain bandwidth product of the CA3140.
C2 R2 R2 C2 1000pF 3 R1 C1000 pF +15V 7 + CA3140 0.1F 6 SUBSTRATE OF CA3019 0.1F -15V R 1 = R2 = R 7 0.1F 7.5k 5 OUTPUT 19VP-P TO 22VP-P THD <0.3%

CA3109 DIODE ARRAY 6

NOTES:

+ OUTPUT

1 f = -----------------------------------------2 R 1 C 1 R 2 C 2 C1 R2 A OSC = 1 + ------ + -----C2 R1 RF A CL = 1 + ------RS
50Hz, R = 100Hz, R = 1kHz, R = 10kHz, R = 30kHz, R =

3.3M 1.6M 160M 16M 5.1M

3.6k 500

RF C1 R1

FIGURE 22. WIEN BRIDGE OSCILLATOR CIRCUIT USING CA3140
Simple Sample-and-Hold System Figure 23 shows a very simple sample-and-hold system using the CA3140 as the readout amplifier for the storage capacitor. The CA3080A serves as both input buffer amplifier and low feed-through transmission switch (see Note 13). System offset nulling is accomplished with the CA3140 via its offset nulling terminals. A typical simulated load of 2k and 30pF is shown in the schematic.
30k STROBE 1N914 +15V 1N914 2k INPUT 5 + 7 + CA0.1 F CA3080A +15V 0.1F 3.5k -15 HOLD 0 SAMPLE
FIGURE 21. BASIC WIEN BRIDGE OSCILLATOR CIRCUIT USING AN OPERATIONAL AMPLIFIER
Oscillator stabilization takes on many forms. It must be precisely set, otherwise the amplitude will either diminish or reach some form of limiting with high levels of distortion. The element, RS, is commonly replaced with some variable resistance element. Thus, through some control means, the value of RS is adjusted to maintain constant oscillator output. A FET channel resistance, a thermistor, a lamp bulb, or other device whose resistance increases as the output amplitude is increased are a few of the elements often utilized. Figure 22 shows another means of stabilizing the oscillator with a zener diode shunting the feedback resistor (RF of Figure 21). As the output signal amplitude increases, the zener diode impedance decreases resulting in more feedback with consequent reduction in gain; thus stabilizing the amplitude of the output signal. Furthermore, this combination of a monolithic zener diode and bridge rectifier circuit tends to provide a zero temperature coefficient for this regulating system. Because this bridge rectifier system has no time constant, i.e., thermal time constant for the lamp bulb, and RC time constant for filters often used in detector networks, there is no lower frequency limit. For example, with 1F polycarbonate capacitors and 22M for the frequency determining network, the operating frequency is 0.007Hz. As the frequency is increased, the output amplitude must be reduced to prevent the output signal from becoming slewrate limited. An output frequency of 180kHz will reach a slew rate of approximately 9V/s when its amplitude is 16VP-P.

For information regarding Intersil Corporation and its products, see www.intersil.com 23

doc1

NOTES: 1. JA is measured with the component mounted on a low effective thermal conductivity test board in free air. See Tech Brief TB379 for details 2. Short circuit may be applied to ground or to either supply.
Electrical Specifications
VSUPPLY = 15V, TA = 25oC TYPICAL VALUES
PARAMETER Input Offset Voltage Adjustment Resistor

SYMBOL

TEST CONDITIONS Typical Value of Resistor Between Terminals 4 and 5 or 4 and 1 to Adjust Max VIO

CA3140 4.7

CA3140A 18

UNITS k

Input Resistance Input Capacitance Output Resistance Equivalent Wideband Input Noise Voltage (See Figure 27) Equivalent Input Noise Voltage (See Figure 35)
RI CI RO eN eN BW = 140kHz, RS = 1M RS = 100 f = 1kHz f = 10kHz

1.4.220

1.4.220 0.4.5 1.4
T pF V nV/Hz nV/Hz mA mA MHz V/s A s % s s
Short Circuit Current to Opposite Supply

IOM+ IOM-

Source Sink
Gain-Bandwidth Product, (See Figures 6, 30) Slew Rate, (See Figure 31) Sink Current From Terminal 8 To Terminal 4 to Swing Output Low Transient Response (See Figure 28)
RL = 2k CL = 100pF RL = 2k CL = 100pF Voltage Follower
Rise Time Overshoot To 1mV To 10mV

0.4.5 1.4

Settling Time at 10VP-P, (See Figure 5)
PARAMETER Input Offset Voltage Input Offset Current Input Current Large Signal Voltage Gain (Note 3) (See Figures 6, 29)
For Equipment Design, at VSUPPLY = 15V, TA = 25oC, Unless Otherwise Specified CA3140 SYMBOL |VIO| |IIO| II AOL MIN TYP 5 0.MAX 50 MIN CA3140A TYP 2 0.MAX 40 UNITS mV pA pA kV/V dB
PARAMETER Common Mode Rejection Ratio (See Figure 34) Common Mode Input Voltage Range (See Figure 8) Power-Supply Rejection Ratio, VIO/VS (See Figure 36) Max Output Voltage (Note 4) (See Figures 2, 8) Supply Current (See Figure 32) Device Dissipation Input Offset Voltage Temperature Drift NOTES: 3. At VO = 26VP-P , +12V, -14V and RL = 2k. 4. At RL = 2k. For Equipment Design, at VSUPPLY = 15V, TA = 25oC, Unless Otherwise Specified (Continued) CA3140 SYMBOL CMRR MIN 70 VICR PSRR -VOM+ VOMI+ PD VIO/T +12 -14 TYP -15.5 to +12.-14.MAX 180 MIN 70 -+12 -14 CA3140A TYP -15.5 to +12.-14.MAX 180 UNITS V/V dB V V/V dB V V mA mW V/oC
For Design Guidance At V+ = 5V, V- = 0V, TA = 25oC TYPICAL VALUES
PARAMETER Input Offset Voltage Input Offset Current Input Current Input Resistance Large Signal Voltage Gain (See Figures 6, 29)
SYMBOL |VIO| |IIO| II RI AOL

CA0.100

CA3140A 2 0.90 -0.5 2.0.3.7 1.200
UNITS mV pA pA T kV/V dB V/V dB V V V/V dB V V mA mA V/s MHz mA mW A
Common Mode Rejection Ratio
Common Mode Input Voltage Range (See Figure 8)

-0.5 2.6

Power Supply Rejection Ratio

PSRR VIO/VS VOM+ VOM-

3 0.3.7 1.200
Maximum Output Voltage (See Figures 2, 8)

Maximum Output Current:

IOM+ I
Slew Rate (See Figure 31) Gain-Bandwidth Product (See Figure 30) Supply Current (See Figure 32) Device Dissipation Sink Current from Terminal 8 to Terminal 4 to Swing Output Low

SR fT I+ PD

CA3140, CA3140A Block Diagram
2mA BIAS CIRCUIT CURRENT SOURCES AND REGULATOR + 3 INPUT 200A 1.6mA 200A A 10,000 C1 12pF STROBE 4 V8 2A 2mA 4mA 7 V+

6 OUTPUT

OFFSET NULL

Schematic Diagram

BIAS CIRCUIT INPUT STAGE SECOND STAGE OUTPUT STAGE DYNAMIC CURRENT SINK 7 V+ D1 Q3 D7 Q2 RR10 1K Q19 RR12 12K Q20 R13 5K

D8 R14 20K

Q17 R1 8K Q8 R8 1K Q 18
6 OUTPUT D2 D3 D4 D5 INVERTING INPUT NON-INVERTING INPUT 2 + 3 RR3 500
Q10 C1 12pF Q14 Q15 Q16 D6 RRQ13

R4 500

R5 500

5 OFFSET NULL

8 STROBE
All resistance values are in ohms.
CA3140, CA3140A Application Information

Circuit Description

As shown in the block diagram, the input terminals may be operated down to 0.5V below the negative supply rail. Two class A amplifier stages provide the voltage gain, and a unique class AB amplifier stage provides the current gain necessary to drive low-impedance loads. A biasing circuit provides control of cascoded constant current flow circuits in the first and second stages. The CA3140 includes an on chip phase compensating capacitor that is sufficient for the unity gain voltage follower configuration. When the CA3140 is operating such that output Terminal 6 is sinking current to the V- bus, transistor Q16 is the current sinking element. Transistor Q16 is mirror connected to D6, R7, with current fed by way of Q21, R12, and Q20. Transistor Q20, in turn, is biased by current flow through R13, zener D8, and R14. The dynamic current sink is controlled by voltage level sensing. For purposes of explanation, it is assumed that output Terminal 6 is quiescently established at the potential midpoint between the V+ and V- supply rails. When output current sinking mode operation is required, the collector potential of transistor Q13 is driven below its quiescent level, thereby causing Q17, Q18 to decrease the output voltage at Terminal 6. Thus, the gate terminal of PMOS transistor Q21 is displaced toward the V- bus, thereby reducing the channel resistance of Q21. As a consequence, there is an incremental increase in current flow through Q20, R12, Q21, D6, R7, and the base of Q16. As a result, Q16 sinks current from Terminal 6 in direct response to the incremental change in output voltage caused by Q18. This sink current flows regardless of load; any excess current is internally supplied by the emitter-follower Q18. Short circuit protection of the output circuit is provided by Q19, which is driven into conduction by the high voltage drop developed across R11 under output short circuit conditions. Under these conditions, the collector of Q19 diverts current from Q4 so as to reduce the base current drive from Q17, thereby limiting current flow in Q18 to the short circuited load terminal.

Input Stage

The schematic diagram consists of a differential input stage using PMOS field-effect transistors (Q9, Q10) working into a mirror pair of bipolar transistors (Q11, Q12) functioning as load resistors together with resistors R2 through R5. The mirror pair transistors also function as a differential-to-single-ended converter to provide base current drive to the second stage bipolar transistor (Q13). Offset nulling, when desired, can be effected with a 10k potentiometer connected across Terminals 1 and 5 and with its slider arm connected to Terminal 4. Cascode-connected bipolar transistors Q2, Q5 are the constant current source for the input stage. The base biasing circuit for the constant current source is described subsequently. The small diodes D3, D4, D5 provide gate oxide protection against high voltage transients, e.g., static electricity.

Bias Circuit

Quiescent current in all stages (except the dynamic current sink) of the CA3140 is dependent upon bias current flow in R1. The function of the bias circuit is to establish and maintain constant current flow through D1, Q6, Q8 and D2. D1 is a diode connected transistor mirror connected in parallel with the base emitter junctions of Q1, Q2, and Q3. D1 may be considered as a current sampling diode that senses the emitter current of Q6 and automatically adjusts the base current of Q6 (via Q1) to maintain a constant current through Q6, Q8, D2. The base currents in Q2, Q3 are also determined by constant current flow D1. Furthermore, current in diode connected transistor Q2 establishes the currents in transistors Q14 and Q15.

Second Stage

Most of the voltage gain in the CA3140 is provided by the second amplifier stage, consisting of bipolar transistor Q13 and its cascode connected load resistance provided by bipolar transistors Q3, Q4. On-chip phase compensation, sufficient for a majority of the applications is provided by C1. Additional Miller-Effect compensation (roll off) can be accomplished, when desired, by simply connecting a small capacitor between Terminals 1 and 8. Terminal 8 is also used to strobe the output stage into quiescence. When terminal 8 is tied to the negative supply rail (Terminal 4) by mechanical or electrical means, the output Terminal 6 swings low, i.e., approximately to Terminal 4 potential.

Typical Applications

Wide dynamic range of input and output characteristics with the most desirable high input impedance characteristics is achieved in the CA3140 by the use of an unique design based upon the PMOS Bipolar process. Input common mode voltage range and output swing capabilities are complementary, allowing operation with the single supply down to 4V. The wide dynamic range of these parameters also means that this device is suitable for many single supply applications, such as, for example, where one input is driven below the potential of Terminal 4 and the phase sense of the output signal must be maintained a most important consideration in comparator applications.

Output Stage

The CA3140 Series circuits employ a broad band output stage that can sink loads to the negative supply to complement the capability of the PMOS input stage when operating near the negative rail. Quiescent current in the emitter-follower cascade circuit (Q17, Q18) is established by transistors (Q14, Q15) whose base currents are mirrored to current flowing through diode D2 in the bias circuit section. When the CA3140 is operating such that output Terminal 6 is sourcing current, transistor Q18 functions as an emitter-follower to source current from the V+ bus (Terminal 7), via D7, R9, and R11. Under these conditions, the collector potential of Q13 is sufficiently high to permit the necessary flow of base current to emitter follower Q17 which, in turn, drives Q18. 5
Output Circuit Considerations
Excellent interfacing with TTL circuitry is easily achieved with a single 6.2V zener diode connected to Terminal 8 as shown in Figure 1. This connection assures that the maximum output signal swing will not go more positive than the zener voltage minus two base-to-emitter voltage drops within the CA3140. These voltages are independent of the operating supply voltage.
V+ 5V TO 36V 8 CA4 6.2V 6 LOGIC SUPPLY 5V TYPICAL TTL GATE
level shifting circuitry usually associated with the 741 series of operational amplifiers. Figure 4 shows some typical configurations. Note that a series resistor, RL, is used in both cases to limit the drive available to the driven device. Moreover, it is recommended that a series diode and shunt diode be used at the thyristor input to prevent large negative transient surges that can appear at the gate of thyristors, from damaging the integrated circuit.

V+ CA10k R 4 6

FIGURE 3A. BASIC
FIGURE 3B. IMPROVED RESOLUTION
FIGURE 3C. SIMPLER IMPROVED RESOLUTION
FIGURE 3. THREE OFFSET VOLTAGE NULLING METHODS
RS LOAD 30V NO LOAD 120VAC CARL MT1 MT7

+HV LOAD 6 RL

CA3140
FIGURE 4. METHODS OF UTILIZING THE VCE(SAT) SINKING CURRENT CAPABILITY OF THE CA3140 SERIES
FOLLOWER +15V 10k CALOAD RESISTANCE (RL) = 2k LOAD CAPACITANCE (CL) = 100pF SUPPLY VOLTAGE: VS = 15V TA = 25oC 6 INPUT VOLTAGE (V) 0 -2 -4 -6 -8 -10 0.1 10mV 1mV 10mV 10 D1 1N914 D2 1N914 1mV 4.99k -15V SETTLING POINT FOLLOWER INVERTING 5k 4 0.1F 5.11k CA100pF 2k 2 10mV 10mV 1mV 1mV 4 0.1F -15V 2k 6 100pF 2k 0.1F SIMULATED LOAD
0.05F INVERTING 5k +15V 7 0.1F SIMULATED LOAD

1.0 SETTLING TIME (s)

FIGURE 5A. WAVEFORM
FIGURE 5B. TEST CIRCUITS FIGURE 5. SETTLING TIME vs INPUT VOLTAGE

Bandwidth and Slew Rate

For those cases where bandwidth reduction is desired, for example, broadband noise reduction, an external capacitor connected between Terminals 1 and 8 can reduce the open loop -3dB bandwidth. The slew rate will, however, also be proportionally reduced by using this additional capacitor. Thus, a 20% reduction in bandwidth by this technique will also reduce the slew rate by about 20%. Figure 5 shows the typical settling time required to reach 1mV or 10mV of the final value for various levels of large signal inputs for the voltage follower and inverting unity gain amplifiers. 7
The exceptionally fast settling time characteristics are largely due to the high combination of high gain and wide bandwidth of the CA3140; as shown in Figure 6.
Input Circuit Considerations
As mentioned previously, the amplifier inputs can be driven below the Terminal 4 potential, but a series current limiting resistor is recommended to limit the maximum input terminal current to less than 1mA to prevent damage to the input protection circuitry. Moreover, some current limiting resistance should be provided between the inverting input and the output when
the CA3140 is used as a unity gain voltage follower. This resistance prevents the possibility of extremely large input signal transients from forcing a signal through the input protection network and directly driving the internal constant current source which could result in positive feedback via the output terminal. A 3.9k resistor is sufficient. The typical input current is on the order of 10pA when the inputs are centered at nominal device dissipation. As the output supplies load current, device dissipation will increase, raising the chip temperature and resulting in increased input current. Figure 7 shows typical input terminal current versus ambient temperature for the CA3140. It is well known that MOSFET devices can exhibit slight changes in characteristics (for example, small changes in
OPEN LOOP PHASE (DEGREES)
input offset voltage) due to the application of large differential input voltages that are sustained over long periods at elevated temperatures. Both applied voltage and temperature accelerate these changes. The process is reversible and offset voltage shifts of the opposite polarity reverse the offset. Figure 9 shows the typical offset voltage change as a function of various stress voltages at the maximum rating of 125oC (for metal can); at lower temperatures (metal can and plastic), for example, at 85oC, this change in voltage is considerably less. In typical linear applications, where the differential voltage is small and symmetrical, these incremental changes are of about the same magnitude as those encountered in an operational amplifier employing a bipolar transistor input stage.

CENTERING -15V 10k 62k 11k 11k 10k 0.1 F EXTERNAL OUTPUT 2 3
7.5k 2M SYMMETRY -15V +15V 100k FROM BUFFER METER DRIVER (OPTIONAL) 39k -15V +
+15V 7 15k 6 7-60 pF HIGH FREQ. SHAPE -15V 51 pF 3 2

+15V 0.1 F

7 + CA3140
HIGH FREQUENCY LEVEL 910k 7-60pF
10k EXTERNAL OUTPUT 6 2.7k -15V TO OUTPUT AMPLIFIER

CA3080A

4 -15V 2k

CA3080 + 4 13k

10k +15V

FREQUENCY ADJUSTMENT

TO SINE WAVE SHAPER
THIS NETWORK IS USED WHEN THE OPTIONAL BUFFER CIRCUIT IS NOT USED

OUTPUT AMPLIFIER

FIGURE 10A. CIRCUIT
Top Trace: Output at junction of 2.7 and 51 resistors; 5V/Div., 500ms/Div. Center Trace: External output of triangular function generator; 2V/Div., 500ms/Div. Bottom Trace: Output of Log generator; 10V/Div., 500ms/Div. FIGURE 10B. FIGURE FUNCTION GENERATOR SWEEPING
METER DRIVER AND BUFFER AMPLIFIER FUNCTION GENERATOR WIDEBAND LINE DRIVER +15V M POWER SUPPLY 15V -15V

SINE WAVE SHAPER

51 GATE DC LEVEL SWEEP ADJUST OFF INT. COARSE RATE VEXT. EXTERNAL INPUT

FINE RATE

SWEEP GENERATOR
1V/Div., 1s/Div. Three tone test signals, highest frequency 0.5MHz. Note the slight asymmetry at the three second/cycle signal. This asymmetry is due to slightly different positive and negative integration from the CA3080A and from the PC board and component leakages at the 100pA level. FIGURE 10C. FUNCTION GENERATOR WITH FIXED FREQUENCIES

SWEEP LENGTH V-

FIGURE 10D. INTERCONNECTIONS
FIGURE 10. FUNCTION GENERATOR
FREQUENCY CALIBRATION MAXIMUM 620k 7 51k 3 + CA3140 3M 4.7k 4 2k +15V 0.1F 12k FREQUENCY 2.4k CALIBRATION MINIMUM 2.5 k 2k METER POSITION ADJUSTMENT 12 3.6k 13
500k FREQUENCY ADJUSTMENT 10k SWEEP IN
TO CA3080A OF FUNCTION CA3080A GENERATOR (FIGURE 10) METER SENSITIVITY ADJUSTMENT 620 1k 200A M METER 14

-15V 0.1F

3 5.1k 2
5.6k 7.5k SUBSTRATE OF CA3019 TO WIDEBAND OUTPUT AMPLIFIER 10k EXTERNAL OUTPUT

50F 25V

2N3053

1N1N914

2.7 2.7
OUTPUT DC LEVEL ADJUSTMENT

+15V 3k -15V 200

2.4pF 2pF

2N4037 -15V

NOMINAL BANDWIDTH = 10MHz tr = 35ns
FIGURE 14. WIDEBAND OUTPUT AMPLIFIER
Power Supplies High input impedance, common mode capability down to the negative supply and high output drive current capability are key factors in the design of wide range output voltage supplies that use a single input voltage to provide a regulated output voltage that can be adjusted from essentially 0V to 24V. Unlike many regulator systems using comparators having a bipolar transistor input stage, a high impedance reference voltage divider from a single supply can be used in connection with the CA3140 (see Figure 15).
A small heat sink VERSAWATT transistor is used as the series pass element in the fold back current system, Figure 17, since dissipation levels will only approach 10W. In this system, the D2201 diode is used for current sampling. Foldback is provided by the 3k and 100k divider network connected to the base of the current sensing transistor. Both regulators provide better than 0.02% load regulation. Because there is constant loop gain at all voltage settings, the
2N6385 CURRENT POWER DARLINGTON LIMITING ADJUST D75 3k 6 2.7k 10F INPUT 2.2k 12 0.01F 4 CA3086 1k 1k 62k HUM AND NOISE OUTPUT <200VRMS (MEASUREMENT BANDWIDTH ~10MHz) LINE REGULATION 0.1%/V LOAD REGULATION (NO LOAD TO FULL LOAD) <0.02% 62k HUM AND NOISE OUTPUT <200VRMS (MEASUREMENT BANDWIDTH ~10MHz) LINE REGULATION 0.1%/V LOAD REGULATION (NO LOAD TO FULL LOAD) <0.02% 4 CA3086 + + CA100k VOLTAGE ADJUST 100k + 3 INPUT 2.2k 0.01F 13 + 1k 82k 2.7k 10F + 8 1k 56pF 180k 2 CA100k VOLTAGE ADJUST 100k + 3 1k 82k 1k 1k 1k 2 2N2102 3
regulation also remains constant. Line regulation is 0.1% per volt. Hum and noise voltage is less than 200V as read with a meter having a 10MHz bandwidth. Figure 18A shows the turn ON and turn OFF characteristics of both regulators. The slow turn on rise is due to the slow rate of rise of the reference voltage. Figure 18B shows the transient response of the regulator with the switching of a 20 load at 20V output.
FOLDBACK CURRENT LIMITER 2N5294 D3 1k 200 OUTPUT 0V TO 25V 25V AT 1A FOLDS BACK TO 40mA

OUTPUT 0.1 24V AT 1A 1

1 100k

3k 2N1k 56pF 180k

5F 50k 14
FIGURE 16. REGULATED POWER SUPPLY
FIGURE 17. REGULATED POWER SUPPLY WITH FOLDBACK CURRENT LIMITING

5V/Div., 1s/Div.

Top Trace: Output Voltage; 200mV/Div., 5s/Div. Bottom Trace: Collector of load switching transistor, load = 1A; 5V/Div., 5s/Div. FIGURE 18B. TRANSIENT RESPONSE
FIGURE 18A. SUPPLY TURN-ON AND TURNOFF CHARACTERISTICS
FIGURE 18. WAVEFORMS OF DYNAMIC CHARACTERISTICS OF POWER SUPPLY CURRENTS SHOWN IN FIGURES 16 AND 17

Tone Control Circuits

High slew rate, wide bandwidth, high output voltage capability and high input impedance are all characteristics required of tone control amplifiers. Two tone control circuits that exploit these characteristics of the CA3140 are shown in Figures 19 and 20. The first circuit, shown in Figure 20, is the Baxandall tone control circuit which provides unity gain at midband and uses standard linear potentiometers. The high input impedance of the CA3140 makes possible the use of lowcost, low-value, small size capacitors, as well as reduced load of the driving stage. Bass treble boost and cut are 15dB at 100Hz and 10kHz, respectively. Full peak-to-peak output is available up to at least 20kHz due to the high slew rate of the CA3140. The amplifier gain is 3dB down from its flat position at 70kHz. Figure 19 shows another tone control circuit with similar boost and cut specifications. The wideband gain of this circuit is equal to the ultimate boost or cut plus one, which in this case is a gain of eleven. For 20dB boost and cut, the input loading of this circuit is essentially equal to the value of the resistance from Terminal No. 3 to ground. A detailed analysis of this circuit is given in An IC Operational Transconductance Amplifier (OTA) With Power Capability by L. Kaplan and H. Wittlinger, IEEE Transactions on Broadcast and Television Receivers, Vol. BTR-18, No. 3, August, 1972.
NOTES: 5. 20dB Flat Position Gain.

5.1 M 2 + CA3140 0.1F 6

FOR SINGLE SUPPLY +30V 2.2M 0.005F
6. 15dB Bass and Treble Boost and Cut at 100Hz and 10kHz, respectively. 7. 25VP-P output at 20kHz. 8. -3dB at 24kHz from 1kHz reference.
FOR DUAL SUPPLIES +15V 0.005F 3 5.1M + CA3140 0.1F 6
BOOST 0.012F 2.2M 0.1 F 18k
TREBLE CUT 200k (LINEAR) 0.001F 100 pF 100pF

4 0.1F -15V

0.022F 2F - + 10k

0.0022F

1M 100k CCW (LOG) BOOST BASS CUT TONE CONTROL NETWORK

TONE CONTROL NETWORK

FIGURE 19. TONE CONTROL CIRCUIT USING CA3130 SERIES (20dB MIDBAND GAIN)
FOR SINGLE SUPPLY BOOST 0.047F BASS CUT (LINEAR) 240k 240k 5M 2.2M 750 pF 750 pF 3 2.2M 0.1 F 2.2 M 2 +32V +15V 7 + CA3140 0.1 F 6 0.047F 4 20pF 5M 51k (LINEAR) BOOST TREBLE CUT TONE CONTROL NETWORK 51k TONE CONTROL NETWORK 7 + CA3140 0.1F 6

FOR DUAL SUPPLIES

4 -15V 0.1F
: 9. 15dB Bass and Treble Boost and Cut at 100Hz and 10kHz, Respectively. 10. 25VP-P Output at 20kHz. 11. -3dB at 70kHz from 1kHz Reference. 12. 0dB Flat Position Gain.
FIGURE 20. BAXANDALL TONE CONTROL CIRCUIT USING CA3140 SERIES

Wien Bridge Oscillator

Another application of the CA3140 that makes excellent use of its high input impedance, high slew rate, and high voltage qualities is the Wien Bridge sine wave oscillator. A basic Wien Bridge oscillator is shown in Figure 21. When R1 = R2 = R and C1 = C2 = C, the frequency equation reduces to the familiar f = 1/(2RC) and the gain required for oscillation, AOSC is equal to 3. Note that if C2 is increased by a factor of four and R2 is reduced by a factor of four, the gain required for oscillation becomes 1.5, thus permitting a potentially higher operating frequency closer to the gain bandwidth product of the CA3140.
C2 R2 R2 C2 1000pF 3 R1 C1000 pF +15V 7 + CA3140 0.1F 6 SUBSTRATE OF CA3019 0.1F -15V R 1 = R2 = R 7 0.1F 7.5k 5 OUTPUT 19VP-P TO 22VP-P THD <0.3%

CA3109 DIODE ARRAY 6

NOTES:

+ OUTPUT

1 f = -----------------------------------------2 R 1 C 1 R 2 C 2 C1 R2 A OSC = 1 + ------ + -----C2 R1 RF A CL = 1 + ------RS
50Hz, R = 100Hz, R = 1kHz, R = 10kHz, R = 30kHz, R =

3.3M 1.6M 160M 16M 5.1M

3.6k 500

RF C1 R1

FIGURE 22. WIEN BRIDGE OSCILLATOR CIRCUIT USING CA3140
Simple Sample-and-Hold System Figure 23 shows a very simple sample-and-hold system using the CA3140 as the readout amplifier for the storage capacitor. The CA3080A serves as both input buffer amplifier and low feed-through transmission switch (see Note 13). System offset nulling is accomplished with the CA3140 via its offset nulling terminals. A typical simulated load of 2k and 30pF is shown in the schematic.
30k STROBE 1N914 +15V 1N914 2k INPUT 5 + 7 + CA0.1 F CA3080A +15V 0.1F 3.5k -15 HOLD 0 SAMPLE
FIGURE 21. BASIC WIEN BRIDGE OSCILLATOR CIRCUIT USING AN OPERATIONAL AMPLIFIER
Oscillator stabilization takes on many forms. It must be precisely set, otherwise the amplitude will either diminish or reach some form of limiting with high levels of distortion. The element, RS, is commonly replaced with some variable resistance element. Thus, through some control means, the value of RS is adjusted to maintain constant oscillator output. A FET channel resistance, a thermistor, a lamp bulb, or other device whose resistance increases as the output amplitude is increased are a few of the elements often utilized. Figure 22 shows another means of stabilizing the oscillator with a zener diode shunting the feedback resistor (RF of Figure 21). As the output signal amplitude increases, the zener diode impedance decreases resulting in more feedback with consequent reduction in gain; thus stabilizing the amplitude of the output signal. Furthermore, this combination of a monolithic zener diode and bridge rectifier circuit tends to provide a zero temperature coefficient for this regulating system. Because this bridge rectifier system has no time constant, i.e., thermal time constant for the lamp bulb, and RC time constant for filters often used in detector networks, there is no lower frequency limit. For example, with 1F polycarbonate capacitors and 22M for the frequency determining network, the operating frequency is 0.007Hz. As the frequency is increased, the output amplitude must be reduced to prevent the output signal from becoming slewrate limited. An output frequency of 180kHz will reach a slew rate of approximately 9V/s when its amplitude is 16VP-P.

-15V 200pF C1

100k -15V 2k

200pF 400

2k 0.1F 30pF SIMULATED LOAD NOT REQUIRED
FIGURE 23. SAMPLE AND HOLD CIRCUIT
In this circuit, the storage compensation capacitance (C1) is only 200pF. Larger value capacitors provide longer hold periods but with slower slew rates. The slew rate is:
dv I ----- = --- = 0.5mA 200pF = 2.5V s dt C NOTE: 13. AN6668 Applications of the CA3080 and CA 3080A High Performance Operational Transconductance Amplifiers.
Pulse droop during the hold interval is 170pA/200pF which is 0.85V/s; (i.e., 170pA/200pF). In this case, 170pA represents the typical leakage current of the CA3080A when strobed off. If C1 were increased to 2000pF, the hold-droop rate will decrease to 0.085V/s, but the slew rate would decrease to 0.25V/s. The parallel diode network connected between Terminal 3 of the CA3080A and Terminal 6 of the CA3140 prevents large input signal feedthrough across the input terminals of the CA3080A to the 200pF storage capacitor when the CA3080A is strobed off. Figure 24 shows dynamic characteristic waveforms of this sample-and-hold system.

Current Amplifier

The low input terminal current needed to drive the CA3140 makes it ideal for use in current amplifier applications such as the one shown in Figure 25 (see Note 14). In this circuit, low current is supplied at the input potential as the power supply to load resistor RL. This load current is increased by the multiplication factor R2/R1, when the load current is monitored by the power supply meter M. Thus, if the load current is 100nA, with values shown, the load current presented to the supply will be 100A; a much easier current to measure in many systems.
R1 10k +15V IL x R2 RM 2 POWER SUPPLY 0.1F 7 + CA0.1F R2 10M IL RL

5 100k 1

Top Trace: Output; 50mV/Div., 200ns/Div. Bottom Trace: Input; 50mV/Div., 200ns/Div.

4.3k -15V

FIGURE 25. BASIC CURRENT AMPLIFIER FOR LOW CURRENT MEASUREMENT SYSTEMS

Note that the input and output voltages are transferred at the same potential and only the output current is multiplied by the scale factor. The dotted components show a method of decoupling the circuit from the effects of high output load capacitance and the potential oscillation in this situation. Essentially, the necessary high frequency feedback is provided by the capacitor with the dotted series resistor providing load decoupling. Full Wave Rectifier Figure 26 shows a single supply, absolute value, ideal fullwave rectifier with associated waveforms. During positive excursions, the input signal is fed through the feedback network directly to the output. Simultaneously, the positive excursion of the input signal also drives the output terminal (No. 6) of the inverting amplifier in a negative going excursion such that the 1N914 diode effectively disconnects the amplifier from the signal path. During a negative going excursion of the input signal, the CA3140 functions as a normal inverting amplifier with a gain equal to -R2/R1. When the equality of the two equations shown in Figure 26 is satisfied, the full wave output is symmetrical.
NOTE: 14. Operational Amplifiers Design and Applications, J. G. Graeme, McGraw-Hill Book Company, page 308, Negative Immittance Converter Circuits.
Top Trace: Output Signal; 5V/Div, 2s/Div. Center Trace: Difference of Input and Output Signals through Tektronix Amplifier 7A13; 5mV/Div., 2s/Div. Bottom Trace: Input Signal; 5V/Div., 2s/Div. LARGE SIGNAL RESPONSE AND SETTLING TIME
SAMPLING RESPONSE Top Trace: Output; 100mV/Div., 500ns/Div. Bottom Trace: Input; 20V/Div., 500ns/Div. FIGURE 24. SAMPLE AND HOLD SYSTEM DYNAMIC CHARACTERISTICS WAVEFORMS
R2 5k +15V 0.1F R10k 3 0.1F 10k INPUT 10k R3 100k OFFSET ADJUST PEAK ADJUST 10k + CA4 1N100pF 2k 7 SIMULATED LOAD +15V

CA3140 + 5

4 0.1F -15V 2k 0.05F
BW (-3dB) = 4.5MHz SR = 9V/s
R2 R3 GAIN = ------ = X = ---------------------------R1 R1 R2 + R3 X+X R 3 = ---------------- R 1 1X R2 5k FOR X = 0.5 -------------- = -----10k R1 0.75 R 3 = 10k ---------- = 15k 0.5

FIGURE 28A. TEST CIRCUIT

20VP-P Input BW (-3dB) = 290kHz, DC Output (Avg) = 3.2V

OUTPUT 0 INPUT 0

Top Trace: Output; 50mV/Div., 200ns/Div. Bottom Trace: Input; 50mV/Div., 200ns/Div. FIGURE 28B. SMALL SIGNAL RESPONSE
FIGURE 26. SINGLE SUPPLY, ABSOLUTE VALUE, IDEAL FULL WAVE RECTIFIER WITH ASSOCIATED WAVEFORMS
+15V RS 3 1M 2 + CA7 0.01F NOISE VOLTAGE OUTPUT

4 0.01F -15V

(Measurement made with Tektronix 7A13 differential amplifier.) Top Trace: Output Signal; 5V/Div., 5s/Div. Center Trace: Difference Signal; 5mV/Div., 5s/Div.
BW (-3dB) = 140kHz TOTAL NOISE VOLTAGE (REFERRED TO INPUT) = 48V (TYP)
Bottom Trace: Input Signal; 5V/Div., 5s/Div. FIGURE 28C. INPUT-OUTPUT DIFFERENCE SIGNAL SHOWING SETTLING TIME FIGURE 28. SPLIT SUPPLY VOLTAGE FOLLOWER TEST CIRCUIT AND ASSOCIATED WAVEFORMS
FIGURE 27. TEST CIRCUIT AMPLIFIER (30dB GAIN) USED FOR WIDEBAND NOISE MEASUREMENT
CA3140, CA3140A Typical Performance Curves
20 GAIN BANDWIDTH PRODUCT (MHz) RL = 2k RL = 2k CL = 100pF 10 25oC 125oC
OPEN-LOOP VOLTAGE GAIN (dB)
TA = -55oC SUPPLY VOLTAGE (V) 25oC 125oC

TA = -55oC

25 SUPPLY VOLTAGE (V)
FIGURE 29. OPEN-LOOP VOLTAGE GAIN vs SUPPLY VOLTAGE AND TEMPERATURE
FIGURE 30. GAIN BANDWIDTH PRODUCT vs SUPPLY VOLTAGE AND TEMPERATURE
RL = QUIESCENT SUPPLY CURRENT (mA) TA = -55oC

RL = 2k CL = 100pF

25oC 125oC 20 SLEW RATE (V/s) SUPPLY VOLTAGE (V) TA = -55oC

25oC 125oC

SUPPLY VOLTAGE (V)
FIGURE 31. SLEW RATE vs SUPPLY VOLTAGE AND TEMPERATURE
SUPPLY VOLTAGE: VS = 15V TA = 25oC 25 OUTPUT SWING (VP-P)
FIGURE 32. QUIESCENT SUPPLY CURRENT vs SUPPLY VOLTAGE AND TEMPERATURE
COMMON-MODE REJECTION RATIO (dB) 120 SUPPLY VOLTAGE: VS = 15V TA = 25oC 101

100K FREQUENCY (Hz)

105 FREQUENCY (Hz)
FIGURE 33. MAXIMUM OUTPUT VOLTAGE SWING vs FREQUENCY
FIGURE 34. COMMON MODE REJECTION RATIO vs FREQUENCY
1000 EQUIVALENT INPUT NOISE VOLTAGE (nV/Hz) SUPPLY VOLTAGE: VS = 15V TA = 25oC POWER SUPPLY REJECTION RATIO (dB) 100 +PSRR

(Continued)

SUPPLY VOLTAGE: VS = 15V TA = 25oC
20 POWER SUPPLY REJECTION RATIO (PSRR) = VIO/VS FREQUENCY (Hz) 106 107

FREQUENCY (Hz)

FIGURE 35. EQUIVALENT INPUT NOISE VOLTAGE vs FREQUENCY
FIGURE 36. POWER SUPPLY REJECTION RATIO vs FREQUENCY
CA3140, CA3140A Metallization Mask Layout

58-66 (1.473-1.676)

0 4-10 (0.102-0.254) 62-70 (1.575-1.778)
Dimensions in parenthesis are in millimeters and are derived from the basic inch dimensions as indicated. Grid graduations are in mils (10-3 inch). The photographs and dimensions represent a chip when it is part of the wafer. When the wafer is cut into chips, the cleavage angles are 57o instead of 90 with respect to the face of the chip. Therefore, the isolated chip is actually 7 mils (0.17mm) larger in both dimensions.

 

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